Transmitter RF power control

ABSTRACT

An electronic system for accurately controlling the RF output power from a power amplifier ( 21 ) is implemented using a multiplier ( 28 ) which multiplies the output power by itself to provide a DC component which is fed to a variable gain amplifier ( 29 ) which provides a controlling signal via a comparator ( 24 ) and an integrator ( 25 ) to the power amplifier. The transfer function between controlling signal and output power is substantially linear, even during dynamic variation of the controlling signal. The system is capable of a large dynamic range, and exhibits constant control loop bandwidth over this range. This is of particular use to TDMA applications with a large dynamic range of levelled output powers because a fixed filter function can be used to shape the transmitted burst.

BACKGROUND OF THE INVENTION

The present invention relates to radio frequency (RF) power control in atransmitter, particularly a transmitter of a cellular telephone handset.

Many radio transmitter systems require accurate control of thetransmitted RF power. This is commonly achieved by a feedback controlsystem in which the difference between the sampled output power from thepower amplifier (PA) and a reference control derives an error signalthat controls the PA gain. This is a common technique because it isusually far easier to implement accurate power measurement with supplyvoltage and temperature variation than it is to maintain PA output powerwithin acceptable limits. Additionally, in many radio transmitterapplications the output power needs to be accurately set to one of anumber of prescribed power levels within some total dynamic range.

TDMA systems usually require the transmitted signal burst to adhere to astrict power versus time template. Thermal heating effects in the PAgenerally cause PA parameters such as gain and/or efficiency to decreasethroughout the burst. A feedback control system can automatically levelthe transmitted burst, provided of course that there is sufficientcontrol loop bandwidth and power output capability. Also, many TDMAsystems require the handset's transmitted power to be smoothly ramped toand from the average burst power. This is commonly specified in order toreduce the amount of adjacent channel interference caused by the powerramp pulse, and/or to limit the power transmitted in adjacent timeslots.

A typical power control loop used in (but not restricted to) mobileradio communications is shown in FIG. 1 of the accompanying drawings. Insuch a control loop, the transmit power P_(trans) output by a poweramplifier 11 (which is fed from a voltage controlled oscillator 10 to anantenna 16) is sampled through a coupler 12 and the value detected in adetector 13 (shown here as a Schottky diode) which outputs a relateddetector voltage V_(det) which is then compared in a comparator 14 witha controllable reference R. The difference signal is integrated in anintegrator 15 and the integrated difference signal ΔS is fed back andused to adjust the gain of the power amplifier 11 in the appropriatedirection in order that the detector voltage V_(trans) moves towards thereference voltage R. This is a classic feedback technique and in thismanner, the controllable reference voltage sets the power output and canbe dynamically controlled to effect the ramp signals required forexample in TDMA system handsets. However, the voltage V_(det) of a diodedetector has a square-law relationship to the transmitted (input) powerP_(trans) and furthermore at low power levels the exponentialcharacteristic of the Schottky diode dominates the response.

In relation to PA output power variance with respect to controlreference, the transfer function of the circuit shown in FIG. 1 isdefined solely by the detector's characteristic. Implementations ofpower detectors are commonly based around either Schottky diodes (asshown) or log-amps. These solutions provide non-linear transferfunctions and also have limited dynamic range, especially in the case ofSchottky diode detectors. This results in the PA power setting withrespect to its control input being non-linear and beyond the control ofthe equipment manufacturer. The transfer function derivative (gradient)therefore varies with power setting and in implementations usingSchottky diodes, the detector usually dominates the overall gradientvariation and implicitly the loop bandwidth variation.

Furthermore, variation of control loop bandwidth with power setting isundesirable and results in a varying dynamic response. In GSM (GlobalSysteme Mobile) systems, which are one form of TDMA system, the rampprofile should ideally be constant with each power setting. Non-lineartransfer functions result in the provision of a unique ramp shape ateach power setting and this demands extra memory. Moreover, the systemis intolerant to unit-to-unit detector and PA variation, as well aspower variation (for example over temperature or due to calibrationerrors).

There is a need therefore to provide a linear control of the RF outputpower from a handset PA and this is preferably achieved by means of afeedback control system.

According to the present invention therefore there is provided RFtransmitter power control system having an oscillator:

-   -   a power amplifier receiving the output of the voltage controlled        amplifier as a first input, having a control input, and        generating an output signal for passing to an antenna;    -   a sampling device for deriving a signal representative of the        output signal;    -   a multiplier for multiplying the representative signal by itself        to provide an output component which is linearly related to the        representative signal;    -   a variable gain amplifier receiving the linear output component        of the multiplier as a first input and having a variable control        input as a control reference and an output;    -   a comparator for comparing the output of the variable gain        amplifier with a reference and producing a difference signal;        and    -   an integrator for integrating the difference signal and        providing it as a control input to the power amplifier.

The way forward to providing a consistent and predictable overall looptransfer function e.g. loop bandwidth, phase margin, is by definition oflinear transfer functions for the individual circuit blocks within theloop. The control loop bandwidth is governed by both the derivatives(rate of change) and absolute gain of the block transfer functions. Alinear transfer function gives a constant derivative.

Several advantages result from the use of the system of the invention,particularly for TDMA mobile communications applications:

-   -   The use of an analogue multiplier to convert the sampled RF        signal to a DC voltage results with a linear relationship        between the transmitted RF voltage and the detected voltage.        This avoids the linearity problems encountered with the use of        Schottky diode detectors or log-amplifiers. The dynamic range of        the multiplier detector is only limited by its large signal        handling and DC offsets or thermal noise at low input signal        powers. The method is identical to a ‘direct conversion        receiver’ architecture and implies the dynamic range is        significantly greater than existing detector solutions.        Furthermore, it is possible to make wideband multiplier circuits        that remove any necessity to incorporate switching for operation        in different frequency bands, or compensate for the slope in        frequency response.    -   The use of a fixed error detection reference combined with a VGA        results in an identical error slope        (dV_((detected))/dV_((transmitted))) for all in-lock RF output        power settings. This results in a constant control loop        bandwidth provided the PA control characteristic is linear. In        practice, this is never the case, but is not the dominant cause        of control loop bandwidth variation in previous typical        implementations.    -   The use of a VGA operating at DC to control the transmitted RF        power can avoid the levelled output RF power drift due to        temperature and voltage variation typically experienced in RF        circuits.

A result of these is that, provided the practical implementation hassufficient control loop bandwidth and dynamic range, the system isfunctionally equivalent to the idealised circuit shown in FIG. 9 below,with the control reference filter equivalent to the closed-loop transferfunction. In bursted TDMA applications, this results in the ability toprofile the transmitted RF signal using a rectangular pulse with heightproportional to transmitted RF voltage and pulse shaping implemented bya fixed filtering function effected either by the closed-loop transferfunction of by an additional dominant filtering function in the controlreference connection. In practical mobile handset applications, thismeans that a single ramp shape can be used together with a scalingmultiplication factor to achieve identical ramp shapes at any transmitRF power. This also means there is no need to use different ramp shapesor incorporate any sensitivity switching between frequency bands ofoperation.

An electronic circuit as proposed is capable of being implemented forpower amplifier operation at very high frequencies as an integratedcircuit or by other suitable components.

The system can of course be used for non-TDMA applications.

One example of an RF power control loop according to the invention willnow be described with reference to the accompanying drawings, in which:

FIG. 2 a block diagram of the control loop;

FIG. 3 shows detail of an idealised multiplier circuit which used in thecontrol loop;

FIG. 4 is a circuit diagram of a conventional Gilbert cell type mixerwhich offers four quadrant multiplication;

FIG. 5 is a circuit diagram of a typical four quadrant multiplier;

FIG. 6 is a circuit diagram the same as FIG. 2, but showing points atwhich signal forms are described in the text below;

FIG. 7 illustrates the operation of the control loop for a typical TDMAapplication;

FIG. 8 is a circuit which allows the use of the RF oscillator at theinput to the power amplifier as the detector's local oscillator;

FIG. 9 illustrates an idealised control loop.

In the circuit shown in FIG. 2, like components to those of FIG. 1 areshown with reference numerals raised by 10. The transmitted signalP_(trans) is sampled in a coupler 22, typically realised using amicrostrip based directional coupler, and is then multiplied by itselfin a mixer circuit or multiplier 28, the multiplying input being derivedfrom the sampled output by an RF limiting amplifier 27. This processresults in the production of sum and difference frequency components ofwhich the difference component is a DC voltage equivalent to thefull-wave rectification of the sampled RF voltage waveform P_(trans).The DC component is fed to a variable gain amplifier 29 which has acontrol reference input V_(control) and an output fed to the poweramplifier's control input via a comparator 24 and an integrator 25.

By way of explanation, the output of the mixer circuit 28 is the productof the two inputs:Output=A ₁·cos(ω·t)·A ₂·cos(ω·t)

The trigonometric identity for cos (a+b) and cos (a−b) can be used toexpress the multiplication as a sum and difference of the frequencyterms (ωt)

${Output} = \left. {{\frac{A_{1} \cdot A_{2}}{2}\mspace{11mu}{\cos\left( {{\omega \cdot t} + {\omega \cdot t}} \right)}} + {\frac{A_{1} \cdot A_{2}}{2}\mspace{11mu}{\cos\left( {{\omega \cdot t} - {\omega \cdot t}} \right)}}}\Rightarrow{{\frac{A_{1} \cdot A_{2}}{2}\mspace{11mu}{\cos\left( {2 \cdot \omega \cdot t} \right)}} + {\frac{A_{1} \cdot A_{2}}{2}\mspace{11mu}{\cos(0)}}} \right.$

Thus the multiplier output has a twice input frequency component and aDC component (cos (0)=1). The twice frequency term can be removed with asuitable filter (not shown).

FIG. 3, which illustrates the idealised mixer circuit or multiplier 28,shows that one of the inputs is an output from the limiting amplifier 27and therefore has an amplitude that is independent of PA output power.Thus the DC component can be represented simply as

${Output} = {\frac{{kA}_{1}}{2}\mspace{11mu}{\cos(0)}\mspace{14mu}{where}\mspace{14mu} k\mspace{14mu}{is}\mspace{14mu} a\mspace{14mu}{{constant}.}}$

In this way the multiplier output can be arranged to be directlyproportional to the output power P_(trans) of the PA, ie. the powerdetection circuit has a linear transfer function.

BRIEF DESCRIPTION OF THE INVENTION

The analysis always holds even if the RF output signal is anglemodulated. Only amplitude modulation is transferred to DC.

In an RFIC (radio frequency integrated circuit) application themultiplier element is likely to be realised with a Gilbert cell typemixer which offers four quadrant multiplication. A generic Gilbert cell,well known in the art, is shown in FIG. 4. The linear differential RFvoltage is applied to RFA and RFB. The differential local oscillatorvoltage is applied to LOA and LOB. As the local oscillator sine waveswings from positive to negative the two differential amplifiers formedby Q3/Q5 and Q4/Q6 steer the currents from Q1 and Q2 to either R1 or R2.This has the effect of reversing the phase of the RF voltage appearingdifferentially at R1 and R2 at a rate equal to the local oscillatorfrequency. This is equivalent to multiplication in the time domain,resulting in sum and difference frequency products appearingdifferentailly at IFA and IFB.

Locating the VGA 29 as shown in FIG. 2, allows it to operate at DC wherethe circuit complexity is significantly reduced from what it would bewere the alternative employed (namely to use a VGA to control RF powerfed into the conventional Schottky diode). This can be done because themixer detector 27, 28 does not suffer the same drawbacks as the diodedetector of the prior art and is able to maintain a well-definedcharacteristic over a wide range of power levels.

The VGA control reference V_(control) sets the output power and can alsobe used to institute the power ramp profile desired for TDMAapplications. This is a departure from a typical power control loopwhere normally the comparator reference determines the output power andramp profile. The VGA gain effectively sets the level of power and thetransmitted output power P_(trans) becomes proportional to the VGA gain.

A typical four quadrant multiplier, as shown in FIG. 5, can be used toimplement the VGA function. See, for example, B. Gilbert “A preciseFour-Quadrant Multiplier with Subnanosecond Response”, IEEE Journal ofSolid-State Circuits, VOL SC-3, No. 4, December 1968. The maindifference to the Gilbert cell is that the local oscillator inputs arelinearised. It produces an output current I₀ that is proportional to theproduct of the input voltages, see the following equation.

$I_{o} = {{\left( {I_{3} + I_{5}} \right) - \left( {I_{4} + I_{6}} \right)} = \frac{2 \cdot {Vx} \cdot {Vy}}{I \cdot {Rx} \cdot {Ry}}}$

In this example, the detector output would be connected across the Yinput and the power control reference across the X input.

The implementation of the comparator and integrator will not be furtherdescribed as their design need not differ fundamentally from those usedin the prior art.

The operation of the control loop for a typical TDMA application isbroken into three different time intervals which are illustrated in FIG.7.

-   -   Power ramping after initialisation.    -   The transition from ramp up to steady state (levelled power).    -   Ramp down.

A diagram of the loop observation points A to F, referenced in FIG. 7,is shown in FIG. 6, which is substantially identical to FIG. 2, but hashad the reference numerals removed for clarity to allow the points A toF to be shown at the desired positions.

Power ramping after initialisation involves the following aspects withinthe circuit:

-   -   1. PA output is assumed to be initially zero. The detector        output voltage is therefore zero.    -   2. A voltage is applied to set the output power (A).    -   3. The VGA inverts and amplifies the detector output (B). The        inversion is required so that the comparator input is the right        sense    -   4. As the negative VGA signal is smaller than the comparator        reference the output of the comparator is positive (C).    -   5. The comparator output is integrated—the resulting output is a        ramp signal (integral of a step input) (D). This is the control        signal for the PA    -   6. The voltage on the control pin increases the output power of        the PA (E).    -   7. The detector output increases as the PA power increases (F).

Transition from ramp up to steady state has the following aspects:

-   -   1. The signal at (B) increases until it is equal to the        reference voltage. The output of the comparator (C) is therefore        zero.    -   2. A zero input to the integrator terminates the ramp signal        (D).    -   3. The output power of the PA is therefore constant (E).    -   4. The loop is now closed

Any residual steady state error is determined by the open loop gain. Ahigh loop gain will lead to a smaller error.

The ramp down process involves:

-   -   1. The voltage of the power setting reference is reduced (A).    -   2. This forces the VGA output to be positive (B) (there is an        inversion).    -   3. The output of the comparator becomes negative (C).    -   4. This negative voltage causes the integrator to ramp down (D).    -   5. Which in turn causes the RF power at the PA output to reduce        (E).    -   6. As the RF power reduces the output from the detector also        reduces (F).

Provided adequate control loop bandwidth exists, the control loop willfollow any profiling applied to the ramping-up and ramping-down process.

DESCRIPTION OF THE INVENTION

The simplest implementation of the power detection part of the controlloop has been depicted in FIG. 2. This implementation avoids anydetected power variation resulting from AM/PM conversion in the poweramplifier 21, provided the limiting amplifier 27 exhibits no AM/PMconversion itself over the desired operating range. In practise, adisadvantage of such an implementation is that the limiting amplifier 27must work over quite a wide range. This means that the limitingamplifier's gain is determined by the minimum required signal to bedetected. Large gain values could lead to issues with RF stability.

Two alternatives to the limiter approach will therefore now bedescribed.

A first alternative is the use of an analogue four quadrant multiplierrather than a conventional Gilbert cell based mixer to square thesampled RF signal. Either the resultant DC signal must then besquare-rooted, or the VGA control signal must be squared in order tomaintain a linear control transfer function. Thus, high gain signallimiting is avoided.

The second alternative is the use of the RF oscillator 20 at the inputto the power amplifier 21 as the detector's local oscillator. Whilstthis completely removes problems associated with limiting amplifierstability or lack of detection sensitivity, it leaves the system proneto detection errors due to AM/PM conversion in the power amplifier 21.However, this problem can be removed by means of the architecture shownin FIG. 8. In this circuit, in phase and quadrature DC components of thetransmitted signal are taken by virtue of two identical Gilbert cellsand a quadrature local oscillator derived from the transmit oscillator.The instantaneous DC components represent the Cartesian vectorrepresentation of the transmitted signal. Any AM/PM conversion in thepower amplifier causes a phase shift to the vector, but the length ofthe vector always remains proportional to the transmitted power. Thevector length is calculated by the Pythagoras theorem.

There still remain some potential practical disadvantages:

-   -   An additional connection to the power control loop is required.    -   There is an increased possibility of unwanted transmitter        oscillator leakage to the antenna.    -   Squaring and square-rooting functions are done by analogue        circuitry.        However, these are outweighed by the ability to provide a linear        power response.

An improvement to the sensitivity of the power control loop can beeffected with the inclusion of automatic DC offset correction.

1. An RF transmitter power control system having an oscillator; a poweramplifier receiving the output of the oscillator as a first input,having a control input, and generating an output signal for passing toan antenna; a sampling device for deriving a signal representative ofthe output signal; a multiplier for multiplying the representativesignal by itself to provide an output component which is linearlyrelated to the representative signal; a variable gain amplifierreceiving the linear output component of the multiplier as a first inputand having a variable control input as a control reference and anoutput; a comparator for comparing the output of the variable gainamplifier with a reference and producing a difference signal; and anintegrator for integrating the difference signal and providing it as acontrol input to the power amplifier.
 2. A system according to claim 1,wherein the oscillator is a voltage controlled oscillator.
 3. A systemaccording to claim 1, wherein the multiplier comprises an analoguemultiplier.
 4. A system according to claim 1, wherein the multipliercomprises a Gilbert cell.
 5. A system according to claim 1, wherein themultiplier has a limiting amplifier input.
 6. A system according toclaim 1, wherein the RF oscillator provides a local oscillator for themultiplier.
 7. A system according to claim 1, wherein the variable gainamplifier comprises a four quadrant multiplier.
 8. An RF transmitterhaving a power control system according to claim
 1. 9. A cellulartelephone handset having an RF transmitter according to claim 8.